Driving systems of AC motor

ABSTRACT

An AC motor control apparatus that is applicable to commonly used PM motors and capable of detecting the magnetic pole position with simple algorithm. The AC motor control apparatus comprises an inverter for applying arbitrary AC power to an AC motor and a controller for sending a control signal to the inverter. The controller comprises a ripple current generator for supplying a ripple current to the AC motor and a magnetic pole position estimator. The magnetic pole position estimator observes at least two current values of the ripple current for both positive and negative sides of the ripple current to estimate the magnetic pole position of the AC motor.

FIELD OF THE INVENTION

The present invention relates to AC motor control apparatus and AC motorsystems that realize the control of motor operation without using asensor to detect an electrical angular position.

BACKGROUND OF THE INVENTION

Prior art of controlling a synchronous motor without detecting anelectrical angular position is disclosed in, for example, JP-A No.2002-78392 (hereinafter referred to as Japanese Patent Document 1) whichrelates to a method for estimating the position of a magnetic polewithin the synchronous motor.

JP-A No. 2001-95215 (hereinafter referred to as Japanese Patent Document2) discloses effects of magnetic saturation occurred locally in a statorof a permanent magnet synchronous motor (PM motor).

The method according to Japanese Patent Document 1 comprises the stepsof applying voltage pulses to the synchronous motor in the directions oftwo axes perpendicular to each other, measuring the amplitudes of thecurrent pulses generated in the directions of both axes, and estimatingthe magnetic pole position, based on the measurements. In this method,by applying approximation to a relation between the generated currentsand the estimated magnetic pole position, compatibility between thenumber of times of applying the voltage pulses and the accuracy of theestimation is achieved.

However, with regard to the change in ripple components of the abovecurrents due to magnetic saturation, Japanese Patent Document 1 makes anassumption as will be described below. FIGS. 14A to 14C show a relationbetween permanent magnet flux φ_(m) and the generated current I_(dc),which is assumed by Japanese Patent Document 1. FIG. 14A shows a dc axisand the direction of permanent magnet flux φ_(m) inside the motor; FIG.14B shows a relation between the current I_(dc) and the primary fluxφ_(Id); and FIG. 14C is a schematic diagram showing the waveform of thecurrent I_(dc). Here, it is supposed that the dc axis along which thevoltage pulses are applied is aligned with the direction of permanentmagnet flux φ_(m) inside the motor as shown in FIG. 14A. When thedirection of the current I_(dc) is aligned with the direction ofpermanent magnet flux φ_(m), the current I_(dc) generates flux in thesame direction as the direction of permanent magnet flux φ_(m), whichacts to accelerate magnetic saturation of the motor core. At this time,inductance L_(ds0) is smaller than inductance L_(d0) that would bemeasured if the direction of the current I_(dc) is opposite to thedirection of permanent magnet flux φ_(m), and the current I_(dc) changesas shown in FIG. 14C.

However, in the case that magnetic saturation occurred locally in thestator of the PM motor has an effect as described in Japanese PatentDocument 2, the assumption by Japanese Patent Document 1 is not alwaystrue, depending on the magnitude of the current I_(dc), and there is apossibility of a major error in estimating the magnetic pole position.The effect of the local magnetic saturation depends on the PM motorstructure and can be reduced relatively by increasing the currentI_(dc), but may be restricted by a controller that drives the motor.

SUMMARY OF THE INVENTION

An object of the invention is to provide an AC motor control apparatusand an AC motor system that are capable of estimating the magnetic poleposition with accuracy.

One feature of the present invention resides in an AC motor controlapparatus comprising a controller which sends a control signal to aninverter which supplies arbitrary AC power to an AC motor, thecontroller comprising a ripple current generator for supplying a ripplecurrent to the AC motor and a magnetic pole position estimator, whereinthe magnetic pole position estimator observes at least two currentvalues of the ripple current for both positive and negative sides of theripple current to estimate the magnetic pole position of the AC motor.

Other features of the present invention are set forth in the appendedclaims.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a system structural diagram according to a preferredembodiment 1 of the present invention.

FIG. 2 is a configuration diagram of a magnetic pole position estimatoraccording to embodiment 1 of the present invention.

FIGS. 3A to 3I show waveform timing charts to explain controlleroperation according to embodiment 1 of the present invention.

FIG. 4A shows an internal view of the motor to control; FIG. 4B shows acurrent-flux relation; and FIG. 4C shows the waveform of the current inorder to explain a relation between magnetic saturation and ripplecurrent if a magnetic pole axis is aligned with the estimated axis.

FIG. 5 is a waveform timing chart to explain controller operationaccording to a preferred embodiment 2 of the present invention.

FIG. 6 is a waveform timing chart to explain controller operationaccording to a preferred embodiment 3 of the present invention.

FIG. 7 is a waveform timing chart to explain controller operationaccording to a preferred embodiment 4 of the present invention.

FIG. 8 is a system structural diagram according to a preferredembodiment 5 of the present invention.

FIG. 9 is a configuration diagram of a magnetic pole position estimatoraccording to embodiment 5 of the present invention.

FIG. 10 is a waveform timing chart to explain controller operationaccording to embodiment 5 of the present invention.

FIG. 11 is a system structural diagram according to a preferredembodiment 6 of the present invention.

FIG. 12 is a configuration diagram of a magnetic pole position estimatoraccording to embodiment 6 of the present invention.

FIG. 13 is a flowchart to explain controller operation according toembodiment 7 of the present invention.

FIG. 14A shows an internal view of the motor to control; FIG. 14B showsa current-flux relation; and FIG. 14C shows the waveform of the currentin order to explain a relation between magnetic saturation and ripplecurrent for a case where a magnetic pole axis is aligned with theestimated axis, which is assumed in prior art.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Embodiment 1

FIG. 1 shows a system structural diagram of a preferred embodiment 1 ofthe present invention. This system has a controller 1 for controllingthe motor, an inverter 2 for driving the motor, and a three-phase ACmotor 3. The controller 1 includes means for changing voltage applied toa dc axis that is an estimated magnetic pole axis inside the motor 3 andmeans for observing current flowing across the motor to estimate themagnetic pole position inside the motor, based on a positive period anda negative period of the ripple components of the observed current.

Concretely, the controller 1 is comprised of a current detector 4 fordetecting current flowing across the motor 3, a dq converter 5 forperforming coordinate conversion from current values into correspondingvalues on dc and qc axes of rotation coordinates the controller, avector controller 6 for controlling the speed or torque of the motor 3,an integrator 7 for integrating electrical angular frequency ω₁ of themotor 3 to calculate an electrical angular position (phase) θ_(dc), a dqinverse converter 8 for performing coordinate conversion from voltagecommands V_(dc)* and V_(qc)* on the dc and qc axes into three-phasevoltage commands, a PWM generator 9 for generating pulses to control theinverter 2, based on a three-phase voltage command, an adder 10 foradding signals, a ripple current generator 11 for applying a voltagesignal to generate a ripple current, a magnetic pole position estimator12 for calculating a position error Δθ (a difference angle betweenactual magnetic pole position and the magnetic pole position that thecontroller assumed) which is a feature of the present invention, a gaincorrector 13 for correcting the electrical angular position θ_(dc),based on the position error Δθ, and an adder 14 for correcting themagnetic pole position inside the controller, based on the magnetic poleposition estimator.

Then, the operation principle of embodiment 1 is discussed. The vectorcontroller 6 performs calculation to control the speed or torque of themotor 3. Three-phase current values detected by the current detector 4are converted by the coordinate converter 5 into corresponding valuesI_(dc) and I_(qc) on the dc and qc axes of the coordinates of rotationinside the controller. The vector controller 6 calculates and outputsvalues of voltages V_(dc0)* and V_(qc0)* to be applied to the motor 3 togive a predetermined value of the I_(dc) component in the directionalong which the magnetic pole of the motor exists and a predeterminedvalue of the I_(qc) component in the direction perpendicular to theabove direction. These voltage commands are converted again intothree-phase AC voltage quantities which are further converted by the PWMgenerator 9 into pulse signals to cause the inverter 2 to performswitching operation. The inverter 2 is driven by the signals from thePWM generator 9 to apply voltages corresponding to the voltage commandscalculated in the controller 1 to the motor 3.

If the phase (position) θ of the magnetic pole of the motor 3 can bedetected directly by a magnetic pole position detector, the detectedthree-phase current values can be coordinate converted, based on thedetected phase. An exciting current component I_(dc) and a torquecurrent component I_(qc) can therefore be obtained. The vectorcontroller 6 controls these two current components separately and has atorque current command and an exciting current command to give desiredspeed and torque of the motor 3. The vector controller 6 changes thevalues of voltages V_(dc0)* and V_(qc0)* to make the detected I_(dc) andI_(qc) values equal to these command values.

As described above, it is necessary to detect the magnetic pole positioninside the motor to perform vector control. The motor driving systemaccording to the present invention is arranged to detect the magneticpole position inside the motor without using the magnetic pole positiondetector (sensor).

Next, the magnetic pole position estimator which is a feature ofembodiment 1 is explained.

FIG. 2 shows a configuration example of the magnetic pole positionestimator 12. FIG. 3 shows waveform timing charts to explain theoperation of magnetic pole position estimator 12 when the ripple currentgenerator 11 gives a voltage command V_(hd)*. Here, it is assumed thatthe three-phase AC motor 3 is now in a stop state, and the dc axis phaseis in a U phase of the stator of the three-phase AC motor 3 in theinitial state, and the inverter output undergoes pulse width modulation.Here, FIG. 3A shows a PWM triangular carrier and it is assumed that onecycle of calculating operation is a half carrier waveform period.

In embodiment 1, the voltage command V_(hd)* shown in FIG. 3B is arectangular waveform voltage whose period is double the PWM triangularcarrier waveform period. At this time, the V_(hd)* voltage is convertedinto three-phase AC voltage quantities, namely, U, V, and W phasevoltage commands V_(hu)*, V_(hv)*, and V_(hw)*, delayed by a halfwaveform period of the PWM triangular carrier corresponding to one cycleof calculating operation of the controller, as shown in FIGS. 3B-2 and3B-3. Furthermore, the U, V and W voltage commands undergoes pulse widthmodulation into U, V, and W phase voltages which are output as shown inFIGS. 3C, 3D, and 3E, respectively.

These voltage outputs generate a ripple current in the three-phase ACmotor. FIG. 3F shows a dc axial current I_(dc) output from thecoordinate converter. Values of the current are detected at timingmarked with dots on the waveform shown in FIG. 3F. At this time, twopair of different absolute values for both the positive polarity and thenegative polarity of the current are obtained, namely, ΔI_(dcp1) andΔI_(dcp2) for a positive half wave and ΔI_(dcn1) and ΔI_(dcn2) for anegative half wave.

Using a delayer 121 and a subtracter 122, the magnetic pole positionestimator calculates a one step difference value ΔI_(dc) for theobtained current values. The one step difference value ΔI_(dc) outputfrom the subtracter 122 is shown in FIG. 3G. Here, an operation delayoccurs by one cycle of detecting the current values. From this one stepdifference value ΔI_(dc), an absolute value calculator 123 obtains itsabsolute value |ΔI_(dc)|. On the other hand, a current polaritycalculator 124 obtains a current polarity signal Sp that indicates thepolarity of the generated dc axial current I_(dc), based on the voltagecommand V_(hd)*. This signal S_(p) should be preferably a rectangularwaveform signal that changes in time with a signal having a delay timeequal to the sum of a delay for one cycle of calculating operation ofthe controller to output the pulse width modulated voltages, a delay forone cycle of detecting the current values to calculate the one stepdifference value ΔI_(dc), and a delay for a quarter waveform period ofthe ripple current shown in FIG. 3F, relative to the voltage commandV_(hd)*. In the instance of embodiment 1, the current polarity signalS_(p) is obtained by delaying the voltage command V_(hd)* by one cycleof detecting the current values and inverting its polarity, as shown inFIG. 3H.

A current variation calculator 125 calculates |ΔI_(dcp)′| and|ΔI_(dcn)′| in the following equation (1) by following a procedure thatwill be described below.

[Equation 1]ΔI _(dcp) ′=ΔI _(dcp2) −ΔI _(dcp1) , ΔI _(dcn) ′=ΔI _(dcn2) −ΔI_(dcn1)  (1)

First, the current polarity signal S_(p) is multiplied to the absolutevalue |ΔI_(dc)|. As a result, a signal including |ΔI_(dcp)′| and|ΔI_(dcn)′| is generated, as shown in FIG. 3I. Then, values areextracted from the waveform of FIG. 3I at timing marked with blacktriangles. Timing marked with the black triangles should be in thecenter between two successive points at which the polarity of thecurrent polarity signal S_(p) changes. The magnetic pole positionestimator 12 calculates the position error Δθ, based on the thusobtained |ΔI_(dcp)′| and |ΔI_(dcn)′|.

Next, the operation principle of estimating the magnetic pole positionin embodiment 1 is discussed. FIG. 4 shows a relation between permanentmagnets inside the motor and the dc axial current I_(dc) generated. FIG.4A shows the dc axis and the direction of permanent magnet flux φ_(m)inside the motor; FIG. 4B is a schematic diagram showing a relationbetween the dc axial current I_(dc) and the primary flux φ_(Id); andFIG. 4C shows the waveform of the dc axial current I_(dc). In thefollowing, it is assumed that the dc axis is aligned with the directionof permanent magnet flux φ_(m) inside the motor as shown in FIG. 4A.

Under the effect of the permanent magnet flux φ_(m), the dc axialcurrent I_(dc) changes asymmetrically with regard to its polarity. Thisis because the permanent magnet flux φ_(m) causes the change ofinductance (L∝dI/dt) with regard to the polarity of the dc axial currentI_(dc). Here, when inductance in the positive direction of the dc axisis denoted by L_(ds0) and inductance in the negative direction of the dcaxis is denoted by L_(d0), there is a relation L_(ds0)<L_(d0).

Meanwhile, if the dc axial current I_(dc) is small, the inductance ismore easily affected by, for example, the stator structure of the motor.Then the inductance becomes L_(ds1) in the positive direction of the dcaxis and L_(ds2) in the negative direction of the dc axis. In FIG. 4B,it is assumed that L_(d0)>L_(ds2) and L_(ds1)>L_(ds0).

When the dc axial current I_(dc) is small (ΔI_(dcn1)<I_(dc)<ΔI_(dcp1)),I_(dc) changes, depending on L_(ds1) in the positive direction of the dcaxis, and L_(ds2) in the negative direction of the dc axis. When the dcaxial current I_(dc) becomes large (I_(dc)<ΔI_(dcn1) orΔI_(dcp1)<I_(dc)), I_(dc) changes, depending on L_(ds0) in the positivedirection of the dc axis, and L_(d0) in the negative direction of the dcaxis.

L_(ds0) and L_(d0) of the inductances are necessary to estimate theposition error Δθ. However, actually detected quantities ΔI_(dcp2) andΔI_(dcn2) of the dc axial current I_(dc) involve the effect of L_(ds0)and L_(ds1) Δ and the effect of L_(d0) and L_(ds2), respectively. Therelation between the waveform of the dc axial current I_(dc) and thedetected current values ΔI_(dcp1), ΔI_(dcn1), ΔI_(dcp2), and ΔI_(dcn2)is shown in FIG. 4C. Then, |ΔI_(dcp)′| and |ΔI_(dcn)′| are obtained,according to the equations of |ΔI_(dcp)′| and |ΔI_(dcn)′|, respectively.|ΔI_(dcp)′| is obtained as a component of the dc axial current I_(dc)which changes depending on L_(ds0). |ΔI_(dcn)′| is obtained as acomponent of the dc axial current I_(dc) which changes depending onL_(d0). Thus, the effect of inductance variation due to the permanentmagnet flux φ_(m) can be extracted as change in the dc axial currentI_(dc).

By the way, with respect to |ΔI_(dcp)′| and |ΔI_(dcn)′| and positionerror Δθ, approximation like, for example, equation (2) below, may beapplied.

[Equation 2]|ΔI _(dcp) ′|−|ΔI _(dcn)′|∝cos(Δθ)  (2)

If this approximation is applied, the range of estimated position errorΔθ is within ±π/2.

According to embodiment 1, the accuracy of estimating the position errorΔθ can be enhanced without being affected by inductance variation due tothe stator structure of the motor.

Embodiment 2

In embodiment 1, timing to detect the current values must be synchronouswith peak and intermediate values of the PWM triangular carrier, asshown in FIG. 3. However, software processing may be complicated togenerate timing synchronous with the intermediate values of the PWMtriangular carrier.

Thus, in a preferred embodiment 2 of the present invention, the overallsystem structure is the same as that shown in FIG. 1 is applied as well,but the voltage command V_(hd)* has a pulse period that is four timesthe PWM triangular carrier waveform period, as is shown in FIG. 5B. Themagnetic pole position estimator 12 operates in the same way as that ofembodiment 1 except that the current polarity signal Sp with polarityopposite to that of the V_(hd)* is obtained. At this time, timing todetect the current values is synchronous with the peak values of the PWMtriangular carrier, as indicated by the dots on the waveform of FIG. 5F.Therefore, according to embodiment 2, the current values can simply bedetected at timing synchronous with the peak values of the PWMtriangular carrier and it is easy to generate timing to detect thecurrent values.

Furthermore, in embodiment 2, if the V_(hd)* pulse period is defined as2n (n is an integer of 2 or greater) times the PWM triangular carrierwaveform period, then the waveform period of the dc axial current I_(dc)is 4n times the half waveform period of the PWM triangular carrier.Thus, n or less than n different absolute values for positive andnegative sides of the dc axial current I_(dc) can be obtained bydetecting the current values at timing synchronous with the peak valuesof the PWM triangular carrier.

Embodiment 3

In a preferred embodiment 3 of the present invention, the overall systemstructure is the same as that shown in FIG. 1, but the voltage commandV_(hd)* with a pulse period that is four times the PWM triangularcarrier waveform period. The voltage V_(hd)* is a stepwise waveform suchthat its amplitude changes in two steps V_(hd1) and V_(hd2), as shown inFIG. 6B. For example, while keeping the average (V_(hd1)+V_(hd2))/2 ofthe amplitude of V_(hd)* constant, by changing the values of V_(hd1) andV_(hd2), the values of ΔI_(dcp1) and ΔI_(dcn1) can be changed, while thepeak values ΔI_(dcp2) and ΔI_(dcn2) of the dc axial current I_(dc)remain constant. The values of ΔI_(dcp1) and ΔI_(dcn1) in embodiment 2of the present invention correspond to those obtained whenV_(hd1)=V_(hd2) in embodiment 3. However, in embodiment 3, a ratiobetween V_(hd1) and V_(hd2) can be selected arbitrarily and, therefore,the values of ΔI_(dcp1) and ΔI_(dcn1) can be set in order not to beaffected by inductance variation due to the stator structure, while thepeak values ΔI_(dcp2) and ΔI_(dcn2) of the dc axial current I_(dc)remain unchanged.

Therefore, according to embodiment 3, the accuracy of estimating theposition error Δθ can be enhanced without increasing the dc axialcurrent I_(dc).

As is the case in embodiment 2, n or less than n different absolutevalues for positive and negative sides of the dc axial current I_(dc)can be obtained by setting the V_(hd)* pulse period 2n (n is an integerof 2 or greater) times the PWM triangular carrier waveform period.

Embodiment 4

In a preferred embodiment 4 of the present invention, the overall systemstructure is the same as that shown in FIG. 1, but the magnetic poleposition estimating operation is divided into two phases: first phaseand second phase as shown in FIGS. 7A to 7I. The voltage command V_(hd)*consists of two rectangular waveform voltages with different amplitudeswhich are applied in two phases, respectively. Specifically, as is shownin FIG. 7B, V_(hd)* with amplitude of V_(hd1)′ is applied in the firstphase and V_(hd)* with amplitude of V_(hd2)′ is applied in the secondphase. The pulse period of V_(hd)* is two times the PWM angular carrierwaveform period in each phase. Timing to detect the values of the dcaxial current I_(dc) is synchronous with the peak values of the PWMtriangular carrier, as indicated by the dots on the waveform of FIG. 7F.The current values ΔI_(dcp1) and ΔI_(dcn1) are to be detected in thefirst phase and the current values ΔI_(dcp2) and ΔI_(dcn2) are to bedetected in the second phase.

In embodiment 4, as is shown in FIG. 71, the product of multiplicationof the current polarity signal S_(p) by the absolute value |ΔI_(dc1) |does not involve information of |ΔI_(dcp)′| and |ΔI_(dcn)′|. However,the maximum values ΔI_(dcp1), ΔI_(dcn1), ΔI_(dcp2), and ΔI_(dcn2) of thedc axial current I_(dc) in each phase are detected. Accordingly, thecurrent variation calculator 125 operates as follows: the currentvariation calculator 125 obtains ΔI_(dcp1) and ΔI_(dcn1) in the firstphase and ΔI_(dcp2) and ΔI_(dcn2) in the second phase as the products ofmultiplication of the current polarity signal S_(p) by the absolutevalue |ΔI_(dcp)′| and, upon the completion of both the first and secondphases, it obtains |ΔI_(dcp)′| and |ΔI_(dcn)′|.

In embodiment 4, V_(hd1)′ and V_(hd2)′ can be set separately. By settingthe values of ΔI_(dcp1) and ΔI_(dcn1) not to be affected by inductancevariation due to the stator structure, while the peak values ΔI_(dcp2)and ΔI_(dcn2) of the dc axial current I_(dc) remain unchanged, theaccuracy of estimating the position error Δθ can be enhanced withoutincreasing the dc axial current I_(dc). ΔI_(dcp1) and ΔI_(dcn1) areequal to the positive and negative peak values of the dc axial currentI_(dc) in the first phase and ΔI_(dcp2) and ΔI_(dcn2) are equal to suchvalues in the second phase. Thus, ΔI_(dcp1), ΔI_(dcn1), ΔI_(dcp2), andΔI_(dcn2) may be used as detected values for the positive and negativesides of the current without calculating the one step difference valueand ΔI_(dcp)′ and ΔI_(dcn)′ can be obtained by equation (1). In thiscase, calculation processing can be simplified.

In embodiments 1 through 4, the voltage command V_(hd)* is a rectangularwaveform voltage that alternates between the positive and negative sidesof one phase axis (dc axis). In this case, it is possible to estimatethe magnetic pole position only in the range of ±π/2 of electricalangles and, an estimation error ±π may exist essentially. Therefore, inorder to realize estimation of the magnetic pole position within ±π ofelectrical angles, it is necessary to use a plurality of theabove-mentioned phase axes.

Embodiment 5

FIG. 8 shows a system structure of a preferred embodiment 5 of thepresent invention. This system differs from the system of FIG. 1 in thatthe ripple current generator 11 outputs a dc axis phase command θ_(dc)_(—) _(ini) simultaneously with the voltage command V_(hd)* to cause thevoltage change and that θ_(dc) _(—) _(ini) is supplied to the input ofthe adder 14. In this structure, the voltage changes in an arbitraryphase of the motor can be achieved by controlling θ_(dc) _(—) _(ini).

FIG. 9 is a configuration diagram of the magnetic pole positionestimator 12 in embodiment 5. FIGS. 10A to 10J show waveform timingcharts to explain the operation of the magnetic pole position estimator12 when the ripple current generator 11 gives the voltage commandV_(hd)*. In this embodiment, as is shown in FIG. 10J, the dc axis phasecommand θ_(dc) _(—) _(ini) to cause the voltage change is changed by 90degrees to cause the voltage change in two directions perpendicular toeach other. In each of two states that θ_(dc) _(—) _(ini) is 0° and thatθ_(dc) _(—) _(ini) is 90°, the voltage command V_(hd)* consists of tworectangular waveform voltages with different amplitudes and with pulseperiod that is double the PWM angular carrier waveform period, which areapplied in sequence, as is the case in embodiment 4. This operation isdivided into four phases: phase d1 during which θ_(dc) _(—) _(ini) is 0°and V_(hd)* with amplitude of V_(hd1)′ is applied; phase d2 during whichθ_(dc) _(—) _(ini) is 0° and V_(hd)* with amplitude of V_(hd2)′ isapplied; phase q1 during which θ_(dc) _(—) _(ini) is 90° and V_(hd)*with amplitude of V_(hd1)′ is applied; and phase q2 during which θ_(dc)_(—) _(ini) is 90° and V_(hd)* with amplitude of V_(hd2)′ is applied.

In the phase d1, at the start of processing, first, the amplitude of thevoltage command V_(hd)* is set at V_(hd1)′ and the dc axis phase commandθ_(dc) _(—) _(ini) to cause the voltage change is set at 0°. Then, theripple current generator 11 applies the V_(hd)* for a preset number ofcycles. In FIGS. 10A to 10J, the number of cycles is set to 2. At thistime, the V_(hd)* undergoes pulse width modulation into U, V, and Wphase voltages which are output as shown in FIGS. 10C, 10D, and 10E,respectively, delayed by a half waveform period of the PWM triangularcarrier corresponding to one cycle of calculating operation of thecontroller. The dc axial current I_(dc) which is generated by thesevoltage outputs is shown in FIG. 10F. Values of the current are detectedat timing marked with dots on the waveform shown in FIG. 10F and currentvalues, ΔI_(dcp1) for positive half waves and ΔI_(dcn1) for negativehalf waves, are obtained. For the thus obtained current values, the onestep difference value ΔI_(dc) calculated through the delayer 121 and thesubtracter 122 is as shown in FIG. 10G. From ΔI_(dc), the absolute valuecalculator 123 obtains its absolute value |ΔI_(dc)|. On the other hand,the current polarity calculator 124 obtains the current polarity signalS_(p), based on the voltage command V_(hd)*. A current amplitudedifference calculator 127 which is a feature of embodiment 5 obtains anintegrated value PF_(d) _(—) _(off) of difference between amplitudeΔI_(dcp1) to the positive polarity and amplitude ΔI_(dcn1) to thenegative polarity of the current I_(dc) generated when the rectangularwaveform voltage with amplitude of V_(hd1)′ is applied in the directionthat θ_(dc) _(—) _(ini)=0°.

[Equation 3]PF _(d) _(—) _(off)=∫(|ΔI _(dcp1) |−|ΔI _(dcn1)|)dt  (3)

After this integrated value is calculated, the processing proceeds tothe phase d2.

In the phase d2, at the start of processing, the amplitude of thevoltage command V_(hd)* is set at V_(hd2)′ and the dc axis phase commandθ_(dc) _(—) _(ini) to cause the voltage change is set at 0°. Then, theripple current generator 11 applies the V_(hd)* for the preset number ofcycles. As is the case of the phase d1, the U, V, and W phase voltagesmodulated from the V_(hd)* are output and the dc axial current I_(dc) isgenerated. The detected current values, ΔI_(dcp2) for positive halfwaves and ΔI_(dcn2) for negative half waves are different from thosedetected in the phase d1. In this phase d2, the current amplitudedifference calculator 127 obtains an integrated value PF_(d) _(—) _(sig)of difference between amplitude ΔI_(dcp2) to the positive polarity andamplitude ΔI_(dcn2) to the negative polarity of the current I_(dc)generated when the rectangular waveform voltage with amplitude of V_(hd)2′ is applied in the direction that θ_(dc) _(—) _(ini)=0°.

[Equation 4]PF _(d) _(—) _(sig)=∫(|ΔI _(dcp2) |−|ΔI _(dcn2)|)dt  (4)

After this integrated value is calculated, the processing proceeds tothe phase q1.

In the phase q1, at the start of processing, the amplitude of thevoltage command V_(hd)* is set at V_(hd1)′ and the dc axis phase commandθ_(dc) _(—) _(ini) to cause the voltage change is set at 90°. Then, theripple current generator 11 applies the V_(hd)* for the preset number ofcycles. As is the case of the phase d1, the U, V, and W phase voltagespulse width modulated from the V_(hd)* are output and the dc axialcurrent I_(dc) is generated. The current values, ΔI_(qcp1) for positivehalf waves and ΔI_(qcn1) for negative half waves are different fromthose detected in the phase d1. In this phase q1, the current amplitudedifference calculator 127 obtains an integrated value PF_(q) _(—) _(off)of difference between amplitude ΔI_(qcp1) to the positive polarity andamplitude ΔI_(qcn1) to the negative polarity of the current I_(dc)generated when the rectangular waveform voltage with amplitude ofV_(hd1)′ is applied in the direction that θ_(dc) _(—) _(ini)=90°.

[Equation 5]PF _(q) _(—) _(off)=∫(|ΔI _(qcp1) |−|ΔI _(qcn1)|)dt  (5)

After this integrated value is calculated, the processing proceeds tothe phase q2.

In the phase q2, at the start of processing, the amplitude of thevoltage command V_(hd)* is set at V_(hd2)′ and the dc axis phase commandθ_(dc) _(—) _(ini) to cause the voltage change is set at 90°. Then, theripple current generator 11 applies the V_(hd)* for the preset number ofcycles. As is the case of the phase d1, the U, V, and W phase voltagesmodulated from the V_(hd)* are output and the dc axial current I_(dc) isgenerated. The detected current values, ΔI_(qcp2) for positive halfwaves and ΔI_(qcn2) for negative half waves are different from thosedetected in the phase d1. In this phase q2, the current amplitudedifference calculator 127 obtains an integrated value PF_(q) _(—) _(sig)of difference between amplitude ΔI_(qcp2) to the positive polarity andamplitude ΔI_(qcn2) to the negative polarity of the current I_(dc)generated when the rectangular waveform voltage with amplitude ofV_(hd2)′ is applied in the direction that θ_(dc) _(—) _(ini)=90°.

[Equation 6]PF _(q) _(—) _(sig)=∫(|ΔI _(qcp2) |−|ΔI _(qcn2)|)dt  (6)

The thus obtained PF_(d) _(—) _(off), PF_(d) _(—) _(sig), PF_(q) _(—)_(off), and PF_(q) _(—) _(sig) have the following relations. First, arelation between PF_(d) _(—) _(off) and PF_(d) _(—) _(sig) is expressedas follows:

[Equation 7]

$\begin{matrix}{{{PF}_{d\_ sig} - {PF}_{d\_ off}} = {{{\int{\left( {{{\Delta\; I_{dcp2}}} - {{\Delta\; I_{dcn2}}}} \right){\mathbb{d}t}}} - \mspace{211mu}{\int{\left( {{{\Delta\; I_{dcp1}}} - {{\Delta\; I_{dcn1}}}} \right){\mathbb{d}t}}}} = {{{\int{\left( {{{\Delta\; I_{dcp2}}} - {{\Delta\; I_{dcp1}}}} \right){\mathbb{d}t}}} - \mspace{211mu}{\int{\left( {{{\Delta\; I_{dcn2}}} - {{\Delta\; I_{dcn1}}}} \right){\mathbb{d}t}}}} = {\int{\left( {{{\Delta\; I_{{dcp}^{\prime}}}} - {{\Delta\; I_{{dcn}^{\prime}}}}} \right){\mathbb{d}t}}}}}} & (7)\end{matrix}$

PF_(d) _(—) _(off) and PF_(d) _(—) _(sig) are PF_(q) _(—) _(off) andPF_(q) _(—) _(sig) when 90 degree rotation and the voltage change occurby the voltage applied by the ripple current generator 11. Therefore, byapplying approximation equation (1) provided in embodiment 1, thefollowing are obtained:

[Equation 8]PF _(d) _(—) _(sig) −PF _(q) _(—) _(off)∝cos(Δθ)  (8)[Equation 9]PF _(q) _(—) _(sig) −PF _(q) _(—) _(off)∝sin(Δθ)  (9)

From these, the magnetic pole position estimator 12 estimates theposition error Δθ, according to the following equation:

[Equation 10]

$\begin{matrix}{{\Delta\;\theta} = {\tan^{- 1}\left( {- \frac{{PF}_{q\_ sig} - {PF}_{q\_ off}}{{PF}_{d\_ sig} - {PF}_{d\_ off}}} \right)}} & (10)\end{matrix}$

Using this equation (10), a position error Δθ can be estimated within±π.

In embodiment 5, the ripple current generator 11 is arranged to apply avoltage to cause the voltage change in two directions perpendicular toeach other and, consequently, estimating the magnetic pole positionwithin ±π of electrical angles can be achieved.

While, in this embodiment, the voltage commands V_(hd)* are applied tocause the voltage change in the order of the phases d1, d2, q1, and q2,the order of these phases may be changed arbitrarily. Even if the orderis changed, magnetic pole position estimation and calculation can beexecuted in the same procedure.

ΔI_(dcp1) and ΔI_(dcn1) are the positive and negative peak values of thedc axial current I_(dc) in the phase d1, and ΔI_(dcp2) and ΔI_(dcn2) aresuch values in the phase d2. ΔI_(qcp1) and ΔI_(qcn1) are such values inthe phase q1, and ΔI_(qcp2) and ΔI_(qcn2) are such values in the phaseq2. Therefore, by using ΔI_(dcp1), ΔI_(dcn1), ΔI_(dcp2), ΔI_(dcn2),ΔI_(qcp1), ΔI_(qcn1), ΔI_(qcp2), and ΔI_(qcn2) as detected values forthe positive and negative sides of the current without calculating theone step difference value and by applying equations (3) to (6),calculation processing can be simplified.

Embodiment 6

FIG. 11 shows a system structure of a preferred embodiment 6 of thepresent invention.

A voltage setting device 15 which is a feature of embodiment 6 may beincorporated in the controller 1 or may be provided outside of thecontroller with communications means for communicating with thecontroller 1. When parameters of the voltage command V_(hd)* such as itspulse amplitude and pulse period are input to the voltage setting device15, the voltage setting device 15 pass these input parameter values tothe ripple current generator 11. The ripple current generator 11 changesthe voltage command V_(hd)* to meet the received input parameter values.This device can realize a function that can change the voltage commandV_(hd)* from the external.

FIG. 12 shows a configuration of the magnetic pole position estimator 12according to embodiment 6. The current amplitude difference calculator127 calculates PF_(d) _(—) _(off), PF_(d) _(—) _(sig), PF_(q) _(—)_(off), and PF_(q) _(—) _(sig) and compares these values withpredetermined values. If all the values of PF_(d) _(—) _(off), PF_(d)_(—) _(sig), PF_(q) _(—) _(off), and PF_(q) _(—) _(sig) are smaller thanthe predetermined values respectively, the current amplitude differencecalculator 127 determines that estimating the position error Δθ will notbe executed properly, because the permanent magnet flux φ_(m) does nothave a significant effect on the dc axial current I_(dc), and outputs anadjustment voltage command signal to the ripple current generator 11.When the adjustment voltage command signal is input to the ripplecurrent generator 11, the ripple current generator 11 increases by apredetermined rate the amplitude of the voltage command V_(hd)* fromnext time. Through this arrangement, such a function can be realizedthat the voltage command V_(hd)* is automatically adjusted so thatestimating the position error Δθ can be executed properly.

Embodiment 7

FIG. 13 shows a flowchart of the operation of the controller 1 accordingto a preferred embodiment 7 of the present invention.

When a system starting command is input to the controller 1, an inverterstarting process 201 is first performed and, upon the completion of theprocess, the controller 1 performs a motor starting command process 202and waits until the motor starting command is input.

When the motor starting command is input to the controller 1, a systemfault diagnosis process 203 which is a feature of embodiment 7 isperformed. The system fault diagnosis process 203 checks for faults suchas a short circuit, grounding, and disconnection in the output circuitof the inverter 2, abnormal conditions such as overvoltage orundervoltage of input voltage, or faults in the converter 1 itself.After the termination of the system fault diagnosis process 203, theprocedure proceeds to a system fault judgment process 204. If a fault isdetected, the procedure proceeds to a system fault recovery process 205.If not, an initial magnetic pole position estimating process 206 isperformed. An initial value of the position error Δθ is estimatedthrough, for example, any one of the methods described in embodiments 1to 5. After this process, the controller starts the motor.

Without the system fault diagnosis process 203 and the system faultjudgment process 204, the motor starts to operate even if any faultoccurs in the system. If, for example, a fault occurs in the currentdetector 4, the initial magnetic pole position estimating process 206cannot estimate the position error Δθ properly and, moreover, thefunction that automatically adjusts the voltage V_(hd)*, described inembodiment 6, may malfunction. According to embodiment 7, thereliability of the method of estimating the magnetic pole positionaccording to the present invention can be enhanced.

As discussed hereinbefore, the present invention is configured to detectthe magnetic pole position and perform vector control by the controllerto control the motor. According to the present invention, the procedurefor detecting the magnetic pole position comprises steps of applying apulsating voltage as the voltage command to cause the voltage changealong the dc axis that is the estimated magnetic pole axis of the motor,observing current flowing across the motor along the dc axis, separatingthe ripple components of the current observed on the dc axis into thepositive and negative sides of the ripple components, detecting two ormore different absolute values of the current for each side andestimating the magnetic pole position inside the motor, based on thethus detected two or more different absolute values of the current.

The above two ore more different absolute values of the current shouldbe detected from one ripple current waveform. Alternatively, it may alsobe preferable to apply voltages with two or more different amplitudes tocause the voltage change along the dc axis and detect one pair ofpositive and negative current values for each voltage.

By moving the dc axis to cause the voltage change in two or moredirections, it is possible to detect the magnetic pole positionincluding polarity.

The controller is provided with the function to set from externally theparameters of the voltage command to cause the voltage change across thedc axis and the function to automatically adjust the voltage commandwithin the controller, so that the magnetic pole position can bedetected even if change is made to the motor.

The controller is provided with the fault detection function to preventmalfunction of the magnetic pole position detecting operation includingthe above automatic voltage adjustment function.

As described hereinbefore, according to the AC motor driving system ofthe present invention, by applying the voltage change to the motor, twoor more different absolute values of ripple components of the currentgenerated by the voltage change are observed in the positive andnegative sides of the current. Based on a current variation ratecalculated from the above current values, the magnetic pole positioninside the motor can be estimated.

The above system takes advantage of ripple current variation due tomagnetic saturation and can remove a portion of the ripple current thatdepends on motor structure. Accordingly, this system is applicableindependent of motor structure and can enhance the accuracy ofestimating the magnetic pole position without increasing the amplitudeof the ripple current.

The controller is provided with the function to set from externally theparameters of the voltage command to cause the voltage change and thefunction to automatically adjust the voltage command within thecontroller. Accordingly, the magnetic pole position can be detected evenif change is made to the motor. The controller is also provided with thefault detection function that can prevent malfunction of the magneticpole position detecting operation including the above automatic voltageadjustment function.

A system comprising an AC motor, an inverter for supplying arbitrary ACpower to the AC motor, and a controller for sending a control signal tothe inverter is referred to as an AC motor system.

The present invention can provide an AC motor control apparatus and anAC motor system that are capable of estimating the magnetic poleposition with accuracy.

1. An AC motor control apparatus comprising a controller which sends a control signal to an inverter which supplies arbitrary AC power to an AC motor, said controller comprising: a ripple current generator for supplying a ripple current to said AC motor; and a magnetic pole position estimator, wherein said ripple current generator outputs a voltage command V_(hd)*, and wherein said magnetic pole position estimator receives said voltage command V_(hd)*, an exciting current component I_(dc) of a motor current, and a torque current component l_(qc) of said motor current as input, and said magnetic pole position estimator outputs a position error Δ⊖ to estimate the magnetic pole position of said AC motor.
 2. The AC motor control apparatus according to claim 1, wherein said magnetic pole position estimator obtains a current variation rate, based on said at least two current values, to estimate the magnetic pole position of said AC motor, based on said current variation rate.
 3. The AC motor control apparatus according to claim 1, wherein said ripple current generator outputs a rectangular waveform voltage as a voltage command.
 4. The AC motor control apparatus according to claim 3, wherein said inverter performs pulse width modulation using a carrier and said rectangular waveform voltage has a pulse period that is four times or greater even integral times a waveform period of the carrier.
 5. The AC motor control apparatus according to claim 1, wherein said ripple current generator outputs a stepwise waveform voltage as the voltage command.
 6. The AC motor control apparatus according to claim 5, wherein said inverter performs pulse width modulation using a carrier and said stepwise waveform voltage has a pulse period that is four times or greater even integral times a waveform period of the carrier.
 7. The AC motor control apparatus according to claim 1, further comprising a current amplitude difference calculator for setting or changing the amplitude of the voltage to be applied.
 8. The AC motor control apparatus according to claim 2, wherein said ripple current generator adjusts the amplitude of a voltage as the voltage command so that the current variation rate of said ripple current falls within a predetermined range.
 9. The AC motor control apparatus according to claim 1, wherein said magnetic pole position estimator for estimating the magnetic pole position inside said motor starts operation after performing a fault detection process to check for a fault in said inverter and said controller.
 10. An AC motor control apparatus comprising a controller which sends a control signal to an inverter which supplies arbitrary AC power to an AC motor, said controller comprising: a ripple current generator for supplying a ripple current to said AC motor; and a magnetic pole position estimator, wherein said ripple current generator outputs rectangular waveform voltages with different amplitudes in sequence as a voltage command V_(hd) * and said magnetic pole position estimator receives said voltage command V_(hd)* an exciting current component I_(dc) of a motor current, and a torque current component l_(qc) of said motor current as input, and said magnetic pole position estimator outputs a position error Δ⊖ to estimate the magnetic pole position of said AC motor.
 11. The AC motor control apparatus according to claim 10, wherein said magnetic pole position estimator obtains a current variation rate, based on said at least one current value, to estimate the magnetic pole position of said AC motor, based on said current variation rate.
 12. The AC motor control apparatus according to claim 10, wherein said inverter performs pulse width modulation using a carrier and said rectangular waveform voltages with different amplitudes have a pulse period that is two times or greater integral times a waveform period of the carrier.
 13. The AC motor control apparatus according to claim 11, wherein observing at least one current value of said ripple current is observing only around positive and negative peaks of said ripple current.
 14. The AC motor control apparatus according to claim 1, wherein said ripple current is applied along two or more arbitrary phase axes.
 15. The AC motor control apparatus according to claim 10, wherein two rectangular waveform voltages with different amplitudes are applied for a predetermined period alternately along a dc axis that is a magnetic pole axis estimated by the controller and along a qc axis perpendicular to the dc axis.
 16. The AC motor control apparatus according to claim 15, wherein said two rectangular waveform voltages with different amplitudes are distinguished as a first rectangular waveform voltage and a second rectangular waveform voltage, a period during which said first rectangular waveform voltage is applied along said dc axis is denoted by d1, a period during which said second rectangular waveform voltage is applied along said dc axis is denoted by d2, a period during which said first rectangular waveform voltage is applied along said qc axis is denoted by q1, a period during which said second rectangular waveform voltage is applied along said qc axis is denoted by q2, and the voltages are applied in sequence of said period d1, said period d2, said period q1, and said period q2 or in sequence of said period d1, said period q1, said period d2, and said period q2.
 17. The AC motor control apparatus according to claim 10, further comprising a current amplitude difference calculator for setting or changing the amplitude of the voltage to be applied.
 18. The AC motor control apparatus according to claim 10, wherein said ripple current generator adjusts the amplitude of a voltage as the voltage command so that the current variation rate of said ripple current falls within a predetermined range.
 19. The AC motor control apparatus according to claim 10, wherein said magnetic pole position estimator to estimate the magnetic pole position inside said motor starts operation after performing a fault detection process to check for a fault in said inverter and said controller.
 20. An AC motor control apparatus comprising a controller which sends a control signal to an inverter which supplies arbitrary AC power to an AC motor, said controller comprising: a ripple current generator for supplying a ripple current to said AC motor; and a magnetic pole position estimator, wherein said ripple current generator outputs a stepwise waveform voltage as the voltage command V_(hd) * or outputs rectangular waveform voltages with different amplitudes in sequence as a voltage command V_(hd) * and said magnetic pole position estimator receives said voltage command V_(hd)*, an exciting current component I_(dc) of motor current, and a torque current component I_(qc) of said motor current as input, and said magnetic pole position estimator outputs a position error Δ⊖ to estimate the magnetic pole position of said AC motor. 